The invention relates to a DC high voltage generator comprising an
AC power supply, an AC-DC rectifier, as well as means for stabilising said DC high
voltage generator against load variations. Such DC high voltage generators are
well known and capable of producing DC voltages ranging from a few kV up to several
MV. In particular, but not limited thereto, such DC high voltage generators are
used to operate linear accelerators in which ions, electrons or other charged
particles are accelerated to high energies. Beside the need for a DC high voltage,
some applications of linear accelerators like electron irradiation, ion implantation
and positron emission tomography (PET), ask for the availability of high output
powers, ranging from a few hundred Watt up to several tens of kW and above. All
these applications have in common that the power from the DC high voltage generator
is used to accelerate a charged particle beam originating from a suitable source.
Today, DC high voltage generators usually apply an all solid state
high frequency (typically 20-200 kHz) switched mode power (SMP) converter that
supplies AC power to an AC-DC rectifier comprising one or more cascade rectifiers
which in turn creates the DC high output voltage. An interface between the converter
and the AC-DC rectifier generally comprises a transformer, a coil and possibly
additional passive electrical components in order to match the converter' s impedance
to that of the cascade rectifier.
In some applications, the AC-DC rectifier(s) is extended with an
electrical resonant circuit to form a high voltage stack. Examples of DC high voltage
generators that apply such a resonant circuit are the "Dynamitron" (see e.g. IEEE
Trans. Nucl Sci. NS-16 (3) (1969),124), the "Cascade transformer high voltage generator"
(US Patent 3,596,167), the "Nested high voltage generator"
5,124,658) and a Cockcroft-Walton high voltage power supply (see e.g. IEEE Trans.
Nucl Sci. NS-16 (3) (1969),117).
However, the sources referred to above are unavoidably susceptible
to sudden discharges in which case the charged particle beam disappears instantaneously
and the needed output power is promptly reduced to nearly zero. Consequently,
these applications, among others, require an optimal transient behaviour of the
DC high voltage generator.
It is well known to those skilled in the art that the output power
of such DC high voltage generators is determined by the duty cycle of the switching
devices in the converter, regardless of the application of one of the described
resonant circuits. During variations in load, the output voltage of the generator
is kept constant by regulating the duty cycle.
A drawback of such an output voltage control is that transient behaviour
depends on the performance of the feedback-loop and consequently overshoot and/or
undershoot during transients are fundamentally unavoidable.
Another drawback of the known high frequency, high power DC generators
is that switching losses present in the converter may become unacceptable if no
appropriate measures are taken. One possibility to eliminate these switching losses
is to operate the converter in zero voltage switching mode (ZVS). ZVS is characterised
in that the turn-on and turn-off of the switching devices is done at moments at
which the voltage across the corresponding switching devices is close to zero.
However, ZVS requires an inductive load to be present at the converter's output.
The performance of DC high voltage generators would therefore greatly
benefit from an electrical design which inherently stabilises the DC output voltage
for optimal transient behaviour and which furthermore enables the switching power
converter to be operated in zero voltage switching mode to virtually eliminate
It is the main purpose of the present invention to create a DC high
voltage generator with optimal transient behaviour. It is also the purpose of the
present invention to realise such an electrical design in a cost-effective manner.
In addition to this, the present invention enables the AC power supply to operate
without swtiching losses and eliminates problems associated with the leakage inductance
and interwinding output capacitance of the transformer.
In order to accomplish these goals, a DC high voltage generator of
the type mentioned in the preamble according to the invention is characterized
in that said stabilising means stabilise said DC high voltage generator by using
at least two electro-magnetically coupled resonant circuits compensating each others
load variation dependency at an operating frequency in such a way that the output
voltage is essentially constant. This provides an inherent stabilisation of the
DC output voltage of the DC high voltage generator during transient conditions.
In a prefered embodiment according to the invention said stabilising
means consist exclusively of passive components, such as inductors and capacitors,
and form an interface between the AC power supply and the AC-DC rectifier. In
general a DC high voltage generator comprises a transformer and according to the
invention at least one passive component may be an integral part of this transformer.
These passive components create a well defined inductive load at
the converter's output, which enables the converter to operate without switching
Thereto said AC power supply, which in general comprises a high frequency
switching power converter, which may comprise BJTs, MOSFETs, IGBTs IGCTs or MCTs
being the switching devices, in a preferred embodiment according to the invention
operates in zero voltage switching mode.
Said AC-DC rectifier can be extended (coupled) with a resonant circuit
to form a high voltage stack, and may be of the "Dynamitron", the "Nested high
voltage generator", the "Cockroft-Walton type voltage multipliers or the " Cascade
transformer high voltage generator" type. In that case the high voltage
stack incorporates one of the at least two electro magnetically coupled resonant
The present invention may be more fully understood from the following
detailed description of the prefered embodiments, reference being made to the accompanying
drawings, in which:
- Figure 1 shows an electrical schematic of the preferred embodiment related
to the present invention.
- Figure 2 shows the same schematic in which essential components are shown in
- Figure 3 is a graph which shows the output voltage Vstack
voltage-current phase difference &phis; of the converter-output as a function of
the output power.
With reference to Figure 1 is shown an electrical schematic of the
preferred embodiment. An AC power supply (1) in the form of a phase-controlled
H-bridge converter comprises four switching devices S1 through S4, and a control
circuit (2). In such a topology, S1 & S2 are known to form one leg of the converter
and are alternatively switched on and off. Similarly, S3 & S4 form the second
leg of the converter. The effective output voltage of the converter that is present
between terminals a&b is controlled by changing the phase between the two
legs of the converter. Any available power switching devices with proper specifications
can be applied in this configuration. Bipolar Junction Transistors (BJTs), Insulated
Gate Bipolar Transistors (IGBTs), Metal Oxide Silicon Field Effect Transistors
(MOSFETs), Mos Controlled Thyristors (MCTs) or Integrated Gate Commutated Thyristors
(IGCTs) are present candidates for the switching devices.
An output transformer (3) is usually applied to match the current-voltage
ratio of the AC power supply (1) to that of The AC-DC rectifier (7). It is readily
understood by those skilled in the art that the switching power converter, which
is characterised in that it has a high power-frequency product, is preferably
operated in zero voltage switching (ZVS) mode, in which switching losses are essentially
reduced to zero. However, ZVS requires that the zero crossings of the converter
output current are lagging the zero crossings of the output voltage, which implies
a dominantly inductive load at the converter output (terminals a&b in figure
Referring again to figure 1, the AC-DC rectifier (7) is extended
by a resonant circuit comprising at least one capacitor (5) Cstack and
one inductor (6) Lstack, to form a high voltage stack (4). By choosing
the operating frequency at or close to the resonance frequency ω0
of the high voltage stack (4), which equals (Cstack&peseta; Lstack)-1/2
for the circuit shown in figure 1, a high AC voltage can be created across terminals
e&f. A AC-DC rectifier (7) connected to these terminals is used to create the
DC high output voltage at terminal (8), which is essentially a fixed multiple
of the peak voltage present at the terminals e&f. It should be noted that the
described high voltage stack (4) is meant for illustration purposes only. Often,
the chosen combination of the resonant circuit and the cascade rectifier (7) will
be more complex and can be based on one of the principles mentioned in one of the
foregoing sections. However, all possible high voltage stacks that can be applied
in conjunction with the preferred embodiment of the present invention have in common
that they operate at or close to a well defined resonance frequency and that they
apply at least one cascade rectifier (7) for the generation of a DC high voltage.
The description of such high voltage stacks is beyond the scope of this writing.
In figure 2 the cascade rectifier (7) is replaced by a load resistor
(9) Rload for the purpose of simplicity only. It shows that the present
invention comprises an inductor (10) Linterface and a capacitor (11)
forming together a second resonant circuit being the interface
(12) between the AC power supply (1) and the high voltage stack (4). With the application
of such an interface (12) the transfer-function H which is defined as:
H = Vstack/Vconverter
can be calculated, with reference to figure 2:
H = H1&peseta;H2
H1 = (ZCinterface//Zstack) / (ZLinterface+ZCinterface//Zstack)
(ZCinterface&peseta;Zstack) / (ZCinterface&peseta;Zstack+ZLinterface&peseta;(ZCinterface+Zstack))
- = Vinterface/Vconverter
- = the voltage at A
- = the voltage at B
- = the voltage at C
H2 = Zout/Zstack
- denotes a parallel connection of two impedances.
- = the impedance of the high voltage stack present at terminals c&d
= Zout+1 /(jωCstack)
- = 2.π.frequency
- = the impedance of Linterface (10)
- = the impedance of Cinterface (11)
= 1 /(jωCinterface)
H = H1 &peseta; H2
- = the impedance formed by the parallel connection of Rload (9) and
Lstack (6) present at the terminals e&f.
jωLstack • Rload / (jωLstack
Under resonance conditions and for relatively low output powers one
Rload >> (JωLstack) or Zout ∼
- = the impedance of Lstack (6)
Zstack ∼ 0, which gives:
H = Zout/Zinterface = ZLstack/ZLinterface
An important feature of the present invention can be seen from equation
1, which shows that under resonance conditions and relatively low output power,
the voltage Vstack at C is essentially a constant. Because the DC output
voltage is a fixed multiple of Vstack, it also implies that the DC output
voltage is, within limits, essentially independent of the output power and therefore
inherently stable during transient conditions.
To illustrate this more strongly, figure 3 gives a graph which shows
the relative output voltage Vef and the voltage-current phase difference
&phis;ab of the converter-output (terminals a&b in fig. 2) as a
function of the output power Pef. Note that
Vef = Vstack
. In the calculations the following input parameters and assumptions were used,
which gives an operating frequency close to 100 kHz.
- = 1.0 milli Henry
- = 2.5 nano Farad
- = 30 milli Henry
- = 85 pico Farad
- = 1 kV RMS
The operating frequency is adjusted to give a constant 20 degrees
capacitive phase at the input terminals of the high voltage stack (terminals c&d
in figure 2). It is readily recognised by those skilled in the art that this assumption
resembles a practical situation in which the operating frequency is set by controlling
the input phase of the high voltage stack.
Referring again to figure 3, it illustrates essential features of
the present invention in that:
- Vef is essentially constant (3% variation in this example) for a
wide range in output powers, which makes the DC output voltage of the high voltage
generator essentially independent of the output power. This in turn implies optimal
- The voltage-current phase difference &phis;ab of the converter-output
is positive under all output power conditions: the phase difference &phis;ab
is lagging in all cases. This implies that, by the application of the interface
circuit, the capacitive load of the high voltage stack (terminals c&d in figure
2) is transferred to an inductive load present at the converter's output (terminals
a&b in figure 2), regardless of the output power of the high voltage stack.
This enables the switching power converter to be operated in zero voltage switching
mode, in which switching losses are essentially zero.
In this example both the resonance frequency of the high voltage
stack as well as that of the interface circuit are chosen to be equal. However,
in a practical design these two resonance frequencies can be chosen sligthly different
to give an optimal adjustment of the transient behaviour as well as the zero voltage
Furthermore, it should be noted that the positions of Linterface
(10) and Cinterface (11) in the interface (12) circuit are not limited
to that shown in figure 2. Although not being the layout of the preferred embodiment,
an interface (12) circuit in which the position of Linterface (10) and
Cinterface (11) are interchanged will function in essentially the same
manner as the interface (12) circuit shown in figure 2. The same holds for the
high voltage stack (4), in which the positions of Lstack (5) and Cstack
(6) can be interchanged if desired, without offending its essential functionality.
It is another important feature of the present invention that it
offers means to eliminate problems associated with the inevitable leakage inductance
and the parasitic interwinding output capacitance present in the high voltage
output transformer (3). Without the presence of the interface (12) circuit, the
leakage inductance and interwinding capacitance will result in reduced available
output power and unwanted oscillations at the output terminals of the transformer
(3), both of which degrade the performance of the entire DC high voltage generator.
For those skilled in the art it is readily understood that with the application
of the present interface (12) circuit such problems are no longer existing because
the leakage inductance and the interwinding capacitance can be thought to be an
integral part of the interface (12) circuit. Hence the inevitable parasitic elements
are made beneficial to the overall system performance.
In fact, it is readily recognised by those skilled in the art that
a proper geometric design of the output transformer (3) will create a leakage inductance
that equals the inductance of Linterface (10) in the
interface (12) circuit. In that case the inductor Linterface (10) will
not be physically present, but will be an integral part of the power converter's
output transformer (3), which reduces costs and complexity.