This invention relates to single phase permanent magnet motors, and
in particular to the sensing of rotor position in such a motor.
Conventional hermetic refrigerator compressors typically use fixed
speed single phase induction motors. Variable speed operation of motors is advantageous
for improving efficiency. Conventional three phase permanent magnet (PM) motors,
such as those being used for heating, ventilating, and air conditioning applications,
are capable of variable speed operation but are more expensive than single phase
PM motors which require fewer power semiconductor switches and associated gate
Refrigeration compressors which are hermetically sealed to prevent
refrigerant leakage have several requirements of their motor drives. Most such
compressors are designed to operate with a preferred direction of rotation due
to the passive lubrication system which usually operates correctly in only one
direction. Furthermore, a three-pin connector has been adopted as an industry standard
for such compressors, so it is advantageous to have a maximum of three wires between
the motor inside the compressor and its controller situated outside the compressor.
The motor additionally should have long term reliability under high temperature
operation (typically 65 °C ambient) and be capable of maintaining output torque
and efficiency at rated speed by maintaining the current in phase with the motor
back electromotive force (EMF) by appropriately phase-advancing the commutation
Single phase PM motors require a suitable current commutation signal
synchronized with the rotor position for proper operation. In most single phase
applications, a Hall-effect position sensor is typically used to detect the rotor
position and thereby control the motor. Such single phase motors having a Hall-effect
sensor, however, generally require a total of five wires: two motor leads and three
leads for the Hall-effect sensor (two too many for the standard sealing connector).
Furthermore, the reliability of such sensors in the compressor environment is uncertain.
In order to avoid the use of a Hall sensor or other rotor position
sensor, various sensorless control schemes have been developed for PM motors. In
three phase PM motors under normal operation, there are times when one phase is
open-circuited and has no current flowing in it. Under such conditions, the terminal
voltage is equal to the back EMF voltage and can thus be sensed directly. Single
phase motors, however, do not have natural intervals where the phase current remains
zero for any length of time, and this approach is therefore not applicable.
For three phase motors, even if the phase current is non-zero, the
back EMF voltage can be calculated by modeling the motor as a resistance, inductance,
and back EMF voltage source, as described by M. Jufer, "Back-EMF Indirect Detection
for Self-Commutation of Synchronous Motors," European Power Electronics Conference,
1987, pp. 1125-29. This technique can also be applied to single phase PM motors
and has the advantage of not requiring any extra sensing leads. However, for single
phase PM motors, it is difficult to provide a controllable preferred direction
of rotation. Thus, the motor can start in either direction, depending on the initial
rotor angular position. Fan and compressor drives generally are designed to operate
in only one direction of rotation, so control over the rotation direction is critical.
Furthermore, the required knowledge of the motor parameters is not always available
and is subject to production and operating variations.
DE2339260 shows a known single phase permanent magnet motor comprising
a rotor, a stator including stator teeth having each one notch and a quadrature
axis winding positioned for generating an output signal representative of rotor
angular position and positioned out-of-phase from a main winding of the stator,
the quadrature axis winding comprising a coil situated in respective notch in the
two stator teeth.
The object of the invention is to provide a single phase permanent
magnet motor having a quadrature axis winding situated in notches where the amplitude
of the output signal is amplified.
According to the invention, there is provided a single phase permanent
magnet motor comprising:
where w is the speed of the rotor, IPEAK is a peak current in the main
winding corresponding to a desired speed and/or torque of the motor, L is the motor
inductance in the main winding, and VS is the voltage across the main
- a rotor;
- a stator including stator teeth, at least two of the stator teeth each having
multiple notches; and
- a quadrature axis winding positioned for generating an output signal representative
of rotor angular position and positioned out-of-phase from a main winding of the
stator, the quadrature axis winding comprising multiple coils connected in series
with each of the coils situated in respective ones of the notches in the at least
two of the stator teeth. The motor may also include a commutation estimator circuit
responsive to the output signal for estimating zero crossings of the back EMF in
the main winding and generating a commutation signal to commutate the main winding
of the stator in advance of the estimated zero crossings of the back EMF by an
angle &thetas; calculated by the following:
&thetas; = L * ω * IPEAK / (Vs)
The invention may best by understood by reference to the following
description taken in conjunction with the accompanying drawings,
where like numerals represent like components, in which:
- Fig. 1 is a flattened sectional top view of a single phase permanent magnet
(PM) motor having a quadrature axis winding for sensing rotor angular position.
- Fig. 1a is a flattened side view of a quadrature axis winding having a multiple
coil structure according to the invention.
- Fig. 2 is a schematic diagram illustrating a direction of forward (preferred)
- Fig. 3 is a graph of voltage, integrated voltage, and phase current versus
time for the diagram of Fig. 2 illustrating an electrical phase lead and commutation
points of the voltage on the quadrature axis winding during low speed operation.
- Fig. 4 is a schematic diagram illustrating a direction of reverse rotation.
- Fig. 5 is a graph of voltage, integrated voltage, and phase current versus
time for the diagram of Fig. 4 illustrating an electrical phase lag and commutation
points of the voltage on the quadrature axis winding during low speed operation.
- Fig. 6 is a graph of integrated voltage and phase current versus time for the
diagram of Fig. 2 illustrating an electrical phase lead and commutation points
of the voltage on the quadrature axis winding during high speed operation.
- Fig. 7 is a block diagram of one embodiment of a single phase PM motor controller
of the present invention.
Fig. 1 is a flattened sectional top view of a single phase permanent
magnet (PM) motor which is not an embodiment of the invention, showing main windings
20 and an additional coil for sensing position (a "quadrature axis winding" 22).
Both the main and the quadrature axis windings are electromagnetically affected
by magnet flux of a rotor 10. However the quadrature winding is not electromagnetically
affected by magnet flux of a stator 12. This allows the quadrature winding to
detect the rotor position without being affected by currents in main winding 20.
Both windings are electromagnetically affected by the magnet flux
of the rotor because the direction of flux to or from the rotor due to each rotor
magnet 14 (represented by flux lines 17 and 19) depends on the polarity of the
rotor magnet adjacent a respective portion of a stator tooth 16. For example, arrows
19 are used to indicate portions of the stator teeth adjacent a magnet having N
(north) type polarity whereas opposing directional arrows 17 are used to indicate
portions of the stator teeth adjacent a magnet having S (south) type polarity.
Therefore, when a tooth is situated such that different portions are adjacent different
polarity magnets, the flux will vary with rotor position. Because of the physical
offset between the main winding and the quadrature axis winding, the magnetic flux
variation and hence the voltage waveforms in the two windings will not be in phase.
The quadrature axis winding is not electromagnetically affected by
the flux from the main winding due to the offset between the windings and because
each portion of a single tooth 16 will transmit stator flux in a common direction
which is not affected by rotor position.
In fig. 1, stator 12 has teeth 16 each having three notches 18, and
quadrature axis winding 22 is wound between the center notches of two teeth and
is therefore about ninety electrical degrees out-of-phase with main winding 20
of the stator. The quadrature axis winding according to the invention has multiple
coils in series (shown in Fig. la) used to increase the magnitude of the induced
quadrature axis winding voltage.
Fig. 1a is a flattened side view of a quadrature axis winding having
a multiple coil structure with leads 126. For example, stator teeth 16 each having
three notches 18 are shown with the quadrature axis winding being wound in three
coil sections 120, 122, and 124 through the teeth with the wound sections coupled
by portions 121 and 123 of the winding. The embodiment of Fig. 1a is useful because
the number of winding turns can be increased over a motor wherein only one coil
section is present and therefore the magnitude of the output signal of the quadrature
axis winding can be amplified.
Fig. 2 is a schematic diagram illustrating a direction of forward
(preferred) rotor rotation, and Fig. 3 is a graph of the main winding back EMF
voltage (Ve), the quadrature winding voltage (Vq), integrated
quadrature winding voltage, and phase current versus time for the diagram of Fig.
2. This graph illustrates the ninety degree electrical phase lead and commutation
points of the voltage on the quadrature axis winding during low speed operation.
The quadrature axis winding voltage, which is directly proportional
to rotor speed and, as discussed above, is out-of-phase with the back EMF of the
main winding, as shown by Fig. 3. Preferably, the degree to which the quadrature
axis winding is out-of-phase ranges from 75 to 105 degrees with an optimum value
being about ninety electrical degrees. However, the invention is expected to work
for any out-of-phase quadrature axis winding capable of providing rotor angular
position control in the manner described below. Whether the quadrature axis winding
electrical phase leads or lags depends upon the direction of rotor rotation, as
The motor controller ideally requires a commutation signal corresponding
to the zero crossings of the main winding back EMF voltage Ve. To obtain
the correspondence, the quadrature axis winding is passed through an integrator
(illustrated as a lead/lag filter, for example, in Fig. 9) which phase retards
the signal. In one embodiment, the phase retard of the integrator is about ninety
degrees. When the quadrature axis winding phase leads by about ninety degrees and
the integrator phase retards the signal by about ninety degrees, the integrated
signal becomes substantially in-phase with the main winding back EMF voltage, as
shown in Fig. 3, and a commutation signal can be obtained by passing the integrated
signal through a comparator (shown in Fig. 9) to detect the zero crossings of
the quadrature axis winding voltage. Each zero crossing represents the initiation
of a change in the polarity of the current in the main winding.
Integrators inherently have gains inversely proportional to their
input frequencies. The voltage induced in the quadrature axis winding is directly
proportional to the motor speed and thus the motor frequency. Therefore, the output
signal of the integrator has an approximately constant amplitude which is independent
of motor speed/frequency; in fact, the output signal of the integrator corresponds
to motor flux. Due to the relatively constant signal amplitude, the zero crossings
can be easily detected over a wide range of speeds. The low frequency cutoff of
the integrator will determine the minimum operating speed.
Fig. 4 is a schematic diagram illustrating a direction of reverse
rotation, and Fig. 5 is a graph of voltage, integrated voltage, and phase current
versus time for the diagram of Fig. 4 illustrating an electrical phase lag and
commutation points of the voltage on the quadrature axis winding.
If rotation of the rotor occurs in a reverse direction (a direction
opposite to the preferred steady-state direction of the motor), as shown in Fig.
5, instead of leading the main winding voltage by ninety degrees, the quadrature
axis winding voltage lags by ninety degrees. Therefore, when the voltage is integrated,
the inherent phase retard of the integrator produces a commutation signal which
is 180 degrees out of phase with the back EMF. This condition produces a braking
torque, as the machine is acting as a generator, which causes the motor to slow
down. Therefore, the motor has only one possible steady state (forward) direction
of rotation - - a feature that is important when driving machines such as fans
The preferred direction of rotation can be easily changed, if desired,
by either inverting the commutation signal or by reversing the polarity of either
quadrature axis winding 22 or main winding 20.
During motor start-up, the motor may transiently rotate in the reverse
direction, depending on the initial rotor position. If the motor does begin to
rotate in the reverse direction, the braking torque will occur, as discussed above,
to stop any such reverse rotation. To improve the motor start-up characteristics
one technique hereinafter referred to as "pre-alignment" is to apply a DC current
to the main winding for a short period on the order of a fraction of a second to
momentarily align the rotor and then, before commutating the current to the main
winding, allow the rotor to fall back to a position in which the starting direction
of the rotor is known.
Another technique hereinafter referred to as "kick-start" is to apply
a DC current pulse of fixed duration ranging from tens of milliseconds to a second
to get the rotor in motion and then switch to the quadrature axis winding signal
for commutation. The commutation signal, which determines the polarity of the
current applied to the motor, is normally obtained from the polarity of the integral
of the quadrature axis signal. When starting, the quadrature coil signal is of
low amplitude and is often noisy. To improve the starting characteristic, the
commutation signal is forced to remain in one state for a fixed duration of tens
of milliseconds to up to a second during the kick-start period. After this period
the polarity of the integral of the quadrature axis voltage is used for commutation
allowing normal operation. The optimum period of the kick-start pulse is determined
by the motor and the load inertia and is best determined experimentally as that
which gives the most reliable starting performance.
These techniques can be applied separately or in combination. Pre-alignment
is designed to ensure a given rotation direction, whereas a kick-start pulse is
expected to improve starting reliability. If both are to be used, pre-alignment
is preferably performed before the kick-start pulse is applied.
Fig. 6 is a graph of integrated voltage and current versus time for
the diagram of Fig. 2 illustrating commutation points of the voltage on the quadrature
axis winding during rated speed operation. During rated speed operation, it is
advantageous to phase advance, i.e., reduce the number of degrees of the phase
retard, in order to maintain the main winding back EMF voltage and the phase current
waveform in phase and thereby obtain maximum output power. The required phase
advance for efficient operation increases from zero degrees (ninety degree phase
retard) at near standstill to typically about 35° (55° phase retard) at rated speed
for single phase PM motors and is a function of speed, bus voltage, and inductance.
Fig. 7 is a block diagram of one embodiment of a single phase PM
motor controller (drive circuit) of the present invention which requires only two
power electronic switches 40 and 42. The quadrature axis winding technique can
also be applied to other single phase controller configurations such as H-bridge
and bifilar configurations. The configuration shown in Fig. 7 has an advantage
for compressor applications as it allows a three-wire motor connection to be more
easily maintained as described below. The power electronic switches may comprise
switches such as MOSFETs or IGBTs, for example.
A 120 V ac line supplies voltage for the motor through a half-bridge
diode/capacitor configuration as follows. A supply lead of the ac line can be coupled
to a first diode 28. A return lead of the ac line can be coupled to a node coupling
a first capacitor 32, a second capacitor 34, and a resistor 36 which are each in
parallel. Second capacitor 34 is coupled at an opposing end to switch 42 and also
to a second diode 30 which in turn directs its output signal to first diode 28.
First diode 28 is coupled to another end of first capacitor 32 as
well as to switch 40. The switches are coupled to the main winding, as well as
to respective gate drives 44 and 46. Resistor 36 is coupled to another side of
the main winding as well as to a hysteresis current controller 50. The main winding
passes a motor current signal Im to the hysteresis current controller
which in turn sends a gate drive signal to the gate drives. Gate drive 46 is coupled
to the hysteresis controller through an inverter 48.
A differential voltage signal Vq from the quadrature axis
winding is directed to a Quadrature Axis Position/Speed Estimator 54 which in turn
sends a commutator signal comm to a commutation block 52 and an estimated speed
signal ω&supand; to a proportional-integral (PI)
speed controller 56. The estimated speed signal is obtained by rectifying the quadrature
axis winding voltage, the mean value of which is proportional to motor speed.
The PI closed loop speed controller uses the estimated speed along with a desired
speed command ω* to determine the current command signal for the commutation
block. The commutation block multiplies the commutation signal (which is indicative
of whether the current signal should be inverted (-1) or unchanged (+1)) and the
current command signal to send a commutated current command signal I* to the hysteresis
The main winding and the quadrature axis winding both require two
connections, so a total of four wires from the motor would normally be required.
To avoid redesigning the existing three-pin connector, one of the connections for
the main winding 20 can be shared with one of the connections from the quadrature
axis winding 22.
Fig. 8 is a circuit diagram illustrating a modified three-pin connection
60 which may be used in the controller embodiment of Fig. 7 for the sharing of
a connection by the main winding 20 and quadrature axis winding 22 in a compressor
case 58. A pseudo "Kelvin" or four-wire connection can be used to minimize the
shared path 20a between a portion of the high current main winding and the low
level quadrature axis winding, and hence reduce the interference.
Fig. 9 is a block diagram of an embodiment of quadrature axis winding
position and speed estimator 54 of the present invention.
A differential amplifier 68 can be used to remove common-mode noise
on the quadrature axis connections. The differential amplifier supplies a signal
to an integrator (shown as lead/lag filter 70) which in turn supplies a phase
retarded signal to zero crossing comparator 72 which provides the commutation signal.
In one embodiment the lead/lag filter has characteristic frequencies of 5 Hz and
175 Hz. The differential amplifier also supplies a signal to a scaling and rectification
block 74 which sends a rectified signal to a lowpass filter 76 for providing the
speed estimation. In one embodiment the lowpass filter has a bandwidth of 24 Hz.
The speed can alternatively be estimated based on the time interval between successive
zero crossings of the phase retarded signal.
The lead/lag filter is designed to phase advance the commutation
signal to maintain the current in phase with the motor back EMF for purposes of
maintaining high torque per ampere and high efficiency as the rotor speed is increased.
With a conventional Hall-effect sensor or back EMF sensing embodiment, phase advancing
the commutation signal requires additional circuitry, such as an analog or digital
phase locked loop, to track the commutation frequency and produce the required
phase advance. Such additional circuitry is not required by the present invention.
Fig. 10 is a circuit diagram of a passive implementation of a first
order lowpass filter which may be used for integrating in the embodiment of Fig.
9. The filter includes a resistor 62 in series with a capacitor 64. The cutoff
frequency of the lowpass filter is designed to be lower than the lowest expected
operating speed, for example, a speed such as 10 rpm. The expected ninety degree
phase shift is obtained at rated speed, for example, at 1000 rpm.
Fig. 11 is a circuit diagram of a passive lead/lag filter which may
be used in the embodiment of Fig. 9. This filter can be fabricated by adding a
single resistor 66 in series with capacitor 64 or a capacitor (not shown) in parallel
with resistor 62 to the embodiment of Fig. 10.
A lead/lag filter can be used to create a phase advance at higher
speeds by designing the filter such that the phase retard reduces as the speed
increases. The passive lead/lag filter can reduce the phase retard from ninety
degrees at medium speeds to a lesser number of degrees (In one embodiment about
55 degrees) at rated speed. This corresponds to 35 degrees of phase advance at
rated speed, which is desirable for efficient motor performance.
Fig. 12 is an analog circuit diagram illustrating one embodiment
for designing the electronic components for the block diagram of Fig. 9.
The quadrature axis winding signal is fed to differential amplifier
68 which comprises a conventional differential amplifier circuit for removing the
common-mode noise on the connections. In one embodiment, the input signals are
fed through resistors 210 and 212 to the input terminals of an operational amplifier
215, a resistor 214 is coupled between a negative input terminal and an output
terminal of operational amplifier 215, and a resistor 216 is coupled between a
positive input terminal of operational amplifier 215 and ground.
The differential amplifier supplies a signal to an active lead/lag
filter 70 which comprises a parallel combination of a capacitor 218 and a resistor
220 coupled to a negative input terminal of an operational amplifier 226, as well
as a parallel combination of a capacitor 222 and a resistor 224 coupled between
the negative input terminal and an output terminal of operational amplifier 226.
Lead/lag filter 70 supplies a phase retarded signal to zero crossing comparator
72 which provides the commutation signal.
The differential amplifier also supplies a signal to a scaler/inverter
74a comprising a resistor 228 coupled to a negative input terminal of an operational
amplifier 232 and a resistor 230 coupled between the negative input terminal and
an output terminal of the operational amplifier.
The inverted signal is then supplied to a rectifier 74b. In one embodiment,
rectifier 74b comprises four resistors 234, 236, 238, and 240 in parallel with
one of the resistors (resistor 240) additionally in parallel with two diodes 242
and 244, one of which is coupled across a negative input terminal and an output
terminal of an operational amplifier 246. Resistors 236 and 238 are further coupled
to a negative input terminal of an operational amplifier 250, with a resistor 248
coupled across the negative input and output terminals of operational amplifier
250 to complete the rectifier.
The output signal of the rectifier is passed through a low pass filter
76 comprising a resistor 252 coupled to a negative input terminal of an operational
amplifier 258, as well as a parallel combination of a capacitor 254 and a resistor
256 coupled between the negative input terminal and an output terminal of operational
amplifier 258. The lowpass filter provides the speed estimation signal.
Referring now to an alternative embodiment, Fig. 13 shows a four-wire
lead count implementation of the invention. As described below, this implementation
of the invention uses zero crossing detection of the quadrature coil signal and
a commutation estimation strategy embodied by a microcomputer and/or adaptive
integrator. Further, the four-wire implementation is driven by an inverter bridge
and performs current regulation by sensing current on a DC link.
As shown in Fig. 13, a motor 300, such as the motor described above,
is for use in driving a rotatable component (not shown). The rotatable component
may be an agitator and/or basket of a laundering apparatus, a fan or blower, or
a compressor as described in, for example, commonly assigned U.S. Patent Nos.
RE 33,655, 5,492,273, 5,418,438, 5,423,192, and 5,376,866, the entire disclosures
of which are incorporated herein by reference. In a preferred embodiment of the
invention, motor 300 is a single phase, electronically commutated motor. It is
to be understood, however, that motor 300 may be any electronically controllable
motor. Such motors may be any electronically controllable motor or dynamoelectric
machine typically powered by an electronic commutating circuit. Such motors include,
for example, external rotor motors (i.e., inside out motors), permanent magnet
motors, single and variable speed motors, selectable speed motors having a plurality
of speeds, and brushless dc motors, including electronically commutated motors,
switched reluctance motors and induction motors. In addition, the motors may be
multiphase motors or single phase motors and, in any case, such motors may have
a single split phase winding or a multi-phase winding. Such motors may also provide
one or more finite, discrete rotor speeds selected by an electrical switch or other
A supply VS provides high voltage DC power to main winding
20 (see Fig. 16) via an inverter bridge 302 (also see Fig. 16). The inverter bridge
302, illustrated as an H-bridge in Fig. 13, includes a plurality of power switches
304, 306, 308, 310 between a positive rail 312 and a negative rail 314. For example,
the power switches 304, 306, 308, 310 may be IGBT's, BJT's or MOSFET's. Inverter
bridge 302 also includes a plurality of flyback diodes 316, 318, 320, 322 corresponding
to switches 304, 306, 308, 310, respectively. Each flyback diode 316, 318, 320,
322 is preferably coupled in an anti-parallel relationship with each switch 304,
306, 308, 310, respectively. By selectively switching power switches 304, 306,
308, 310 to connect supply VS to winding 20, inverter bridge 302 provides
power to winding 20 in at least one preselected sequence for commutating main
winding 20. In this embodiment, main winding 20 is the direct or torque-producing
coil of motor 300 (see Fig. 16). It is to be understood that supply VS
may also provide power to operate the various other circuits in the system.
According to one embodiment of the invention, a commutation estimator
circuit 324 (see Fig. 15) generates motor control signals, or commutation signals,
for commutating winding 20. In a preferred embodiment, the commutation estimator
circuit 324 is embodied by a microcontroller or microcomputer which executes routines
for determining optimum commutation instances as a function of the desired speed
and/or torque of motor 300. As such, commutation estimator circuit operates as
a state machine. In response to the commutation signals, motor 300 produces a peak
current that corresponds to the load torque demand. The current in winding 20 in
turn produces an electromagnetic field for rotating rotor 10 of motor 300. By
matching torque load with produced torque, motor 300 operates at a desired torque
The commutation signals preferably include a series of pulse width
modulated cycles, wherein each cycle causes a corresponding switching event of
power switches 304, 306, 308, 310. Winding 20 of motor 300 is adapted to be commutated
in at least one preselected sequence and power switches 304, 306, 308, 310 selectively
provide power to winding 20 in the preselected sequence. By regulating current
and, thus, torque, in motor 300, the load and motor loss demand torque may be
matched so that motor 300 achieves the desired speed. In the alternative, it is
contemplated that a voltage regulated control strategy, rather than a current regulated
strategy, may be implemented for controlling speed and/or torque of motor 300.
In a preferred embodiment, inverter bridge 302 operates from a single
commutating signal which selects either switches 304 and 310 or switches 306 and
308 depending on the position of rotor 10. In this embodiment, only one of the
active switches (e.g., switch 308 or 310) is involved in current regulation (pulse
width modulating) at any given time. By performing pulse width modulation, inverter
bridge 302 preferably provides a peak current to winding 20 that corresponds to
the desired speed and/or torque of motor 300. As an example of the normal motoring
operation of motor 300, a set of gate drives (not shown) enable a pair of switches,
such as switches 304 and 310, in response to a commutation signal generated by
commutation estimator circuit 324. Commutation estimator circuit 324 causes the
pair of switches 304, 310 to be enabled wherein one of the two switches (e.g.,
switch 310) performs pulse width modulation while the other remains in its on
state for the entire commutation interval as commanded by the commutation logic.
The polarity of the motor back EMF during this time interval is counter to the
supply voltage Vs
such that the driving electromotive force developing
current in motor 300 is the supply VS minus the back EMF. Commonly
assigned U.S. Patent No. 4,757,603, for example, shows a pulse width modulation
control for a motor.
Referring further to Fig. 13, commutation estimator circuit 324 receives
signals via line 326 representative of the position of rotor 10 (see Fig. 8). For
example, quadrature winding 23, illustrated in Fig. 1g, provides rotor position
feedback. As shown in Fig. 1g, quadrature winding 23 is positioned along the center
of each stator tooth 16 and wound from tooth to tooth along stator 12. Notch 18
of each tooth 16 retains winding 23 in place. Due to its position in the center
of teeth 16, quadrature winding 23 provides for cancellation of flux originating
from torque producing currents circulating in main winding 20. In other words,
the magnetic flux of adjacent poles cancel so that the magnetic flux of stator
12 does not affect the voltage induced in quadrature winding 23. Further, since
quadrature winding 23 is wound along the entire stator 12 and positioned out-of-phase
from winding 20, the voltage induced in the quadrature coil is a phase-shifted
signature of the voltage induced in main winding 20. The quadrature signal also
accounts for all asymmetries rotor 10 may have because it is essentially an average
signal for all of the stator teeth 16. The averaging effect of the quadrature coil
associated with pole alignment on the motor magnets has a beneficial effect on
the bidirectionality of the motor (i.e., its ability to produce the same torque
in both directions) in the presence of unevenness of the magnet arcs due to manufacturing
The magnitude of the quadrature voltage signal depends on the speed
of rotor 10, the stack length of the laminations comprising stator 12, the magnet
strength and length of rotor magnet 14 (see Fig. 1), and the number of poles.
The shape of the quadrature signal, which is a true signature of the back EMF induced
in main winding 20, is influenced by rotor skew, magnet asymmetries and lamination
design. Other means including Hall sensors, slotted disks with opto interrupters,
and the like may also provide rotor position feedback for motor 300 instead of
or in addition to quadrature winding 23. For example, Hall sensors provide a rotor
feedback signal which is typically in phase with the back EMF in main winding
20. However, as described above, Hall sensors require more connectors than quadrature
winding 23 which is undesirable in certain applications. To optimize commutation
of motor 300, such position feedback means typically require precise control of
positioning tolerances and accurate measurements.
In a preferred embodiment of the present invention, quadrature winding
23 (see Fig. 16) comprises a multiple-turn coil. For example, quadrature winding
23 comprises a six-turn winding coil for twelve stator poles. In the case of a
multiple-turn coil, the voltage induced in quadrature winding 23 is the summation
of the individual voltages induced in each single turn of winding 23. By summing
the individual voltages, the quadrature winding signal accommodates for differences
from coil to coil in the voltages sensed in individual coils due to, for example,
As described above, the zero crossings of the back EMF in quadrature
winding 23 provide information regarding the zero crossings of the back EMF in
winding 20. Commutation estimator circuit 324 (see Fig. 15) preferably determines
the position of rotor 10 as a function of the zero crossings of the quadrature
winding signal and generates commutation signals in response thereto. Torque production
in motor 300 is then determined by the product of the current and the back EMF
in winding 20. In order to sustain positive torque, commutation estimator circuit
324 commutates winding 20 at an angular distance prior to the zero crossing of
the back EMF wave in the direction that will oppose the voltage energizing it.
With the correct angular distance, current in winding 20 reaches zero at the time
that the back EMF also reaches zero.
A shunt resistor, current transformer, Hall-effect current sensor,
integrated current sensor or other sensor or circuit known in the art may be used
to sense the winding or motoring current of motor 300. As illustrated in Fig. 13,
inverter bridge 302 includes a single resistive shunt RSHUNT in the
negative rail 314. Only the motor current flows through the shunt resistor RSHUNT
when power is being exchanged from supply VS to motor 300 and vice versa.
Fig. 14 illustrates exemplary waveforms of the signals processed
by commutation estimator circuit 324 with respect to time. Fig. 14(a) shows the
idealized voltage in quadrature winding 23 (i.e., Vq). Fig. 14(a) also
shows the idealized back EMF waveform (i.e., Ve or -Ve depending
on the direction of rotation) in main winding 20. Observed from quadrature winding
23, the voltage induced in the torque producing coil (i.e., main winding 20) is
shown for different directions of rotation. As described above, quadrature axis
winding 23 is preferably positioned approximately 90° out-of-phase from main winding
20. Thus, the phase difference between the two signals is approximately 90°. As
an example, the 90° phase difference is indicated in Fig. 14(a) as the difference
between a zero crossing 328 on the Vq waveform and zero crossings 330
on the ±Ve waveforms. Fig. 14(b) shows the digital representations of
the quadrature and direct coil signals, referred to as Zq and ±Ze.
For example, the digital representation Zq of the quadrature winding
signal Vq is obtained by detecting the zero crossing of the waveform
using a comparator (see Fig. 15).
Since back EMF signals are generated only when rotor 10 is moving,
position information is not available when motor 300 is at standstill. Thus, in
a preferred embodiment, motor 300 initially operates according to a starting algorithm.
For example, inverter bridge 302 (see Fig. 16) applies current to motor winding
20 by turning on one pair of switches on opposite legs of inverter bridge 302.
In the case of split capacitor topology, such as shown in Fig. 7, one switch is
turned on. As a result of the current applied to winding 20, motor 300 generates
torque in either direction of rotation and rotor 10 starts moving. As soon as rotor
10 moves, voltage is induced in quadrature winding 23 and the digital representation
of this signal (i.e., Zq) is available for processing.
Once rotor 10 is moving, however, commutation is preferably synchronized
to the quadrature signal Vq. Fig. 15 shows a block diagram of a preferred
circuit for processing the quadrature winding signal Vq during start-up
and running conditions. According to this embodiment of the invention, a zero
crossing comparator 332 first generates the signal Zq at line 334. After
rotor 10 begins moving during start-up, an integrator 336 (see Fig. 15) integrates
the digital quadrature winding signal Zq and outputs a signal INT via
line 338 representative of the integration. Following integration, a comparator
340 compares the integrated signal INT to a reference level VREF. As
a result of the comparison, comparator 340 provides a commutation signal via line
342 for commutating motor 300. On the other hand, commutation estimator circuit
324 outputs commutation signals via line 344 to commutate winding 20 during running
A selector 346 outputs either the signal via line 342 from the integrator
336 and comparator 340 or the signal via line 344 from commutation estimator circuit
324 to commutate motor winding 20. In a preferred embodiment, the selector 346
comprises a switch responsive to the speed of rotor 10 for selecting between the
commutation signal via line 342 from comparator 340 when the speed of rotor 10
is less than a threshold speed (e.g., 120 rpm) or the signal via line 344 from
commutation estimator circuit 324 when the speed of rotor 10 reaches the threshold
speed. In other words, after reaching the threshold speed, the start-up procedure
concludes and selector 346 switches control to commutation estimator circuit 324.
In the event rotor 10 has not reached the threshold starting speed, inverter bridge
302 may be force-commutated at a certain speed (e.g., 40 rpm).
As described above, commutation estimator circuit 324 may be embodied
as a microcontroller. In a preferred embodiment, the microcontroller monitors the
speed of rotor 10 by measuring the time interval between commutation instances.
As an example, for a twelve-pole motor with 180° conduction intervals, twelve
commutation instances occur for every mechanical revolution of the rotor. The number
of commutations per mechanical revolution varies with the number of rotor poles.
The length of the conduction interval could be less than 180° in certain applications
(see Fig. 18). In this manner, the microcontroller is able to calculate the speed
of motor 300 based on the time interval between commutations. In the alternative,
it is to be understood that various other speed sensors or circuits may be used
for detecting the speed of rotor 10.
Advantageously, the circuit of Fig. 15 provides a high level of noise
immunity at low speeds where the amplitude of the quadrature signal Vq
is likely to be small and subject to noise. As speed of rotor 10 increases, the
amplitude of the quadrature signal Vq increases and provides well defined
transitions at the zero crossings of the waveform. According to the invention,
the circuit of Fig. 15 may be implemented in hardware using analog integration
and a phase lock loop circuit. Alternatively, a microcontroller executes routines
to perform the integration of the quadrature winding signal or to perform another
starting algorithm as well as the estimation of the commutation instances. Fig.
16 illustrates a preferred motor drive circuit in schematic form which incorporates
the features of the circuit of Fig. 15.
As shown in Fig. 16, a motor drive system according to the invention
includes a user interface, or input and output interface I/O, which cooperates
with a nonvolatile memory EE for providing system control signals to commutation
estimator circuit 324, embodied as a microcontroller. In the illustrated embodiment,
commutation estimator circuit 324 provides a commutation signal that includes two
drive control signals, referred to as CMM1 and CMM2, via line 348 to a set of logic
gates 350. The logic gates 350 output the commutation signals in the form of gate
drive signals for electronically controlling a pair of upper gate drives 352, 354
and a pair of lower gate drives 356, 358. In turn, the gate drives 352, 354, 356,
358 provide enough signal conditioning to switch power switches 304, 306, 308,
310, respectively. In addition to providing voltage signals shifted from, for example,
5 volts to 15 volts for driving the power switches, gate drives 352, 354, 356,
358 also condition the signals provided from commutation estimator circuit 324
via line 348 for optimal operation of power switches 304, 306, 308, 310.
An AC to DC voltage converter 360 provides supply voltage VS
(shown as +DC link and -DC link in Fig. 16) via a DC link (i.e., rails 312 and
314 of inverter bridge 302) to power switches 304, 306, 308, 310 for commutating
a winding 20 of motor 300. The AC to DC voltage converter 360 also provides low
voltage power (shown as low voltage sources VC and VD in
Fig. 16) to operate commutation estimator circuit 324 and other circuitry of the
motor drive system.
As described above, matching torque load with produced torque causes
motor 300 to operate at a desired torque or speed. According to the invention,
the motor drive system of Fig. 16 includes a current regulator circuit 362 for
causing motor 300 to produce a peak current that matches the load torque demand
as a function of a peak regulated current reference signal IREF. The current regulator
circuit 362 converts the digital IREF signal to analog and compares it to the
sensed current in the DC link. Then, as a function of a pulse width modulation
frequency signal fPWM, current regulator circuit 362 outputs a PWM signal to the
logic gates 350 which is a function of the peak regulated current. In this manner,
current regulator circuit 362, in cooperation with commutation estimator circuit
324, regulates current in motor 300.
In a preferred embodiment of the invention, the motor drive system
also includes an overcurrent detector circuit 364 for independently comparing the
sensed current to a maximum current reference shown as ITRIP*. The maximum current
reference is, for example, 20-50% greater than the peak regulated current level.
Preferably, the maximum current reference is fixed according to the power limitations
of power switches 304, 306, 308, 310 and/or motor 300. According to the invention,
overcurrent detector circuit 364 compares the sensed current in the DC link to
the maximum current reference and generates an overcurrent signal ITRIP when the
sensed current exceeds the maximum current reference. In turn, commutation estimator
circuit 324 receives the ITRIP signal and sets the state machine of the microcontroller
accordingly to disable power switches 304, 306, 308, 310 and disconnect winding
20 from the supply VS.
Fig. 16 also illustrates a current zero crossing comparator circuit
366 for generating a signal Izero
representative of the zero crossings
of the current sensed in the DC link. A signal IZREF provides a reference for the
zero current detector Izero. Preferably, its level is set slightly
greater than zero.
Referring again to Fig. 14, Fig. 14(c) illustrates the integral of
signal Zq (i.e., the digital representation of the quadrature coil voltage),
INT, on integrator 336 via line 338. In this approach, the amplitude of Zq
is independent of motor speed. Fig. 14(d) illustrates an exemplary commutation
signal CMM, which is a function of the integrated signal INT, for causing commutation
in motor 300. In this case, the advance angle is zero and integrator 336 provides
90° delay from the quadrature coil signal. Commutation instances 368 coincide
with the zero crossings of the torque producing coil voltage, indicated at reference
character 330 on the Ve and -Ve
waveform, for example. This
allows maximum accelerating torque at low speeds. Integrator 336, as shown in Fig.
15, is used from start-up to a threshold speed.
Further to the invention, commutation estimator circuit 324 determines
preferred commutation instances for optimizing the performance of motor 300. As
stated above, motor winding 20 must be energized at a proper instant of time relative
to the generated back EMF to develop optimum motoring torque in motor 300. Fig.
14(d) shows commutation instances 368 which coincide with the zero crossings of
the torque producing coil, or direct coil, voltage (Ze, -Ze).
However, due to the inductive nature of motor 300, which is driven by a voltage
source inverter such as inverter bridge 302, the motor current takes a finite time
to reach a desired current level and to decay toward zero from a determined current
level. Thus, motor current must be commutated in anticipation of the transition
of the back EMF waveform in order for motor current to actually cross zero at the
time the motor back EMF also crosses zero.
Commutation estimator circuit 324 preferably provides further optimization
of motor 300 by causing the actual commutation instances to occur before motor
back EMF crosses zero. By commutating motor 300 before motor back EMF crosses
zero, the motor inductive current is allowed to decay while the back EMF also approaches
zero. This allows the fundamental components of motor current to be in phase with
the generated back EMF which maximizes torque production. In other words, motor
300 preferably achieves a unity power factor.
According to a preferred embodiment of the present invention, commutation
estimator circuit 324 estimates a commutation angle in terms of the inductance
of motor 300, the supply voltage VS, the commanded motor current, and
the speed of rotor 10. Commutation estimator circuit 324 further tunes the estimated
angle for maximum torque production by observing the zero crossings of the motor
current and synchronizing the zero crossings of the current with the zero crossings
of the back EMF. This approach is valid for steady state operation.
An open loop equation for determining the commutation angle according
to the invention may be derived using a simplified motor voltage equation. Neglecting
the winding resistance and assuming that the motor back EMF in the vicinity of
a commutation instance is close to zero, the following equation is the voltage
across motor winding 20:
VS = L di / (dt)
Expressing the above equation in terms of a rotating speed, ω,
and solving for an angular distance:
VS = L di / (d&thetas;) * d&thetas; / (dt)
d&thetas; = L * d&thetas; / (dt)*di / (VS)
&thetas; = L * ω * IPEAK / (VS)
where &thetas; is the angle it takes for the current to decay towards zero, ω
is the rotor speed, IPEAK is the peak commanded current before the commutation,
L is the motor inductance, and VS is the voltage across winding 20.
This simplified equation for &thetas;-sets a first order approximation for the
advance angle but does not require extensive computation power as the complete
equation would. It is to be understood that the above equation may be modified
so as to calculate an advance time interval rather than an angle because of the
relationship between motor speed and angular distance.
Another way to effect an advance angle for commutating motor winding
20 is by adaptive delaying of the integrator signal INT. A variable delay circuit
(not shown) following integrator 332 accommodates the start and running commutation
angle requirements over the entire speed range.
Fig. 17 illustrates exemplary waveforms for motor 300 with respect
to time in which the advance angle &thetas; is calculated according to the equation
described above. Fig. 17(a) shows the idealized direct back EMF waveform Ve
relative to the idealized quadrature back EMF waveform Vq. Fig. 17(b)
shows the digital signals Zq and Ze representing the zero
crossings of the quadrature and back EMF signals, respectively. According to a
preferred embodiment of the invention, the microcomputer embodying commutation
estimator circuit 324 generates the digital signal Ze by first measuring
the interval between zero crossings of the quadrature winding signal. In the illustrated
embodiment, quadrature winding 23 is preferably 90° out-of-phase with main winding
20 which usually conducts in 180° intervals. For this reason, the zero crossings
of the motor back EMF may be calculated by dividing the interval between zero
crossings of the quadrature signal by two. Fig. 17(b) indicates a quadrature signal
zero crossing by reference character 372 followed by another quadrature signal
zero crossing indicated by reference character 374. The microcomputer first measures
the interval T between the zero crossings 372, 374 and then estimates that the
next zero crossing of the motor back, indicated by reference character 376, will
follow by an interval T/2.
Fig. 17(c) shows an exemplary commutation signal CMM generated by
commutation estimator circuit 324. As indicated by reference characters 378 and
380, commutation preferably occurs at the angle &thetas; in anticipation of the
back EMF zero crossings 382 and 376, respectively. Fig. 17(d) illustrates the
motor current IM relative to the commutation and back EMF signals and
Fig. 17(e) illustrates the current detected at the DC link shunt RSHUNT
(see Fig. 16), i.e., Ishunt. In this instance, the current in the shunt
is the current being exchanged between supply VS and motor winding 20.
The intervals in which Ishunt is zero correspond to pulse width modulation
off intervals when motor current IM is decaying while circulating in
motor winding 20 and inverter bridge 302. When winding 20 is commutated, Ishunt
has an opposite polarity (e.g., as shown at reference character 384) indicating
the current flow from winding 20 to supply VS.
As shown in Fig. 17(d), beginning at reference character 386, the
motor current IM starts to decay toward zero immediately following the
commutation instance 378 and crosses zero at reference character 388. Likewise,
beginning at reference character 390, the motor current IM starts to
decay toward zero immediately following the commutation instance 380 and crosses
zero at reference character 392. Advantageously, commutation estimator circuit
324 provides further tuning of the estimated angle for maximum torque production
by observing the zero crossings of the motor current during steady state operation
and synchronizing them with the zero crossings of the back EMF.
Comparator circuit 366 is used to detect the zero crossings of shunt
current Ishunt to determine the zero crossings of the motor current
IM. The waveform of Fig. 17(f) represents the output Izero
of such a comparator. As described above, commutation estimator circuit 324 preferably
causes commutation in motor 300 to occur so that the fundamental components of
motor current are in phase with the generated back EMF to maximize torque production.
By comparing the signal Izero to the estimated zero crossing signal
Ze, commutation estimator circuit 324 determines the relative displacement
between the back EMF and the current zero crossings. Although the commutation
instance 378 does not yet provide optimum torque, as indicated by the displacement
δ, the information of the relative displacement between the back EMF and
the current zero crossing provided by signal Izero allows the commutation
angle to be adjusted to minimize this displacement. For example, the transition
of Izero shown at reference character 394, which corresponds to the
current zero crossing 388, occurs before the back EMF zero crossing 382, indicating
a negative displacement δ. The angular difference between the time inductive
current decays to zero and the time the back EMF crosses zero must be minimized
for optimal commutation. During steady state operation, commutation estimator circuit
324 observes this difference and adjusts, or fine tunes, the advance angle to
minimize the displacement δ. In this instance, commutation estimator circuit
324 decreases the advance angle (e.g., &thetas;1). As a result of the
adjustment, the transition of Izero shown at reference character 396,
which corresponds to the current zero crossing 392, coincides with the voltage
zero crossing 376. Thus, torque production in motor 300 is optimized. When angular
displacement δ is greater than a minimum angle and inverter bridge 302 is
enabled for 180° conduction intervals, negative torque is produced by motor 300.
This negative torque reduces the motoring torque and could generate audible noise.
Alternatively, the circuit of Fig. 16, which includes a commutation
signal comprised of two control signals (i.e., CMM1 and CMM2), can be commanded
to operate at conduction intervals less than 180°. Motor conduction angles less
than 180° reduces torque production and can be used for torque control whenever
optimal operation is not required. With this scheme, an interval of zero current
"dead time" is inserted after the motor current is commutated and has reached zero.
The use of dead time is beneficial in preventing the generation of negative torque
when rapid acceleration or deceleration is required and the estimation of the back
EMF is subject to errors.
Fig. 18 illustrates exemplary waveforms for motor 300 with respect
to time in which the conduction intervals are less than 180°. Fig. 18(a) shows
the idealized direct back EMF waveform Ve relative to the idealized
quadrature back EMF waveform Vq. Fig. 18(b) shows the digital signals
Zq and Ze
representing the zero crossings of the quadrature
and back EMF signals, respectively. Fig. 18(c) shows a pair of exemplary control
signals CMM1 and CMM2 constituting the commutation signal generated by commutation
estimator circuit 324. As indicated by reference characters 398 and 400, commutation
preferably occurs at the angle &thetas;1 and the angle &thetas;2
in anticipation of back EMF zero crossings 402 and 404, respectively. Fig. 18(d)
illustrates the motor current IM
relative to the commutation and back
EMF signals and Fig. 18(e) illustrates the current detected at the DC link shunt
(see Fig. 16), i.e., Ishunt. As shown in Figs. 18(d)
and 18(e), commutating main winding for less than 180° causes a dead time in the
current. The dead time is shown by, for example, an off interval 406. As shown
in Fig. 18(d), beginning at reference character 408, the motor current IM
starts to decay toward zero immediately following the commutation instance 398.
After reaching zero, the current is off for the off interval 406 until the next
commutation at reference character 410. Likewise, beginning at reference character
412, the motor current IM starts to decay toward zero immediately following
the commutation instance 400. Again, after reaching zero, the current is off for
the off interval 406 until the next commutation at reference character 414. Thus,
as shown in Figs. 18(d) and 18(e), the current waveforms have intervals of no
current due to the conduction intervals being less than 180°. Note that the motor
current decays to zero and remains at zero twice every electrical cycle.